Detecting the number of transmit antennas in wireless communication systems

ABSTRACT

To detect the number of transmit antennas, a fast Fourier transform operation is performed on the received samples of the transmitted long training symbols of a preamble. Next, each of the Fourier transformed results is multiplied with the reference frequency-domain representation of the long training symbol so as to remove the effect of the symbols and to maintain the channel information. Next, inverse Fourier transform or least squares operations is performed on the multiplied values to compute channel impulse response. The number of shifted impulse response in the channel impulse response represents the detected number of transmit antennas. Packets containing preambles of the present invention may be received by extended devices as well as by legacy receivers that are not configured to receive and interpret these preambles. The training symbols may be cyclically-shifted and transmitted on different transmit antennas.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a Continuation of U.S. application Ser. No.11/139,925, filed May 27, 2005, entitled “DETECTING THE NUMBER OFTRANSMIT ANTENNAS IN WIRELESS COMMUNICATION SYSTEMS” which is aNon-Provisional of U.S. Provisional Application No. 60/575,608, filedMay 27, 2004, entitled “MODIFIED PREAMBLE STRUCTURE FOR IEEE 820.11AEXTENSIONS AND DETECTING THE NUMBER OF TRANSMIT ATTENAS IN MIMO OR MISOCOMMUNICATION SYSTEMS”, the content of each is hereby incorporated byreference in its entirety.

The present application is also related to co-pending U.S. applicationSer. No. 10/820,440, filed Apr. 5, 2004, entitled “MODIFIED PREAMBLESTRUCTURE FOR IEEE 802.11A EXTENSIONS TO ALLOW FOR COEXISTENCE ANDINTEROPERABILITY BETWEEN 802.11A DEVICES AND HIGHER DATA RATE, MIMO OROTHERWISE EXTENDED DEVICES”, the contents of which is incorporatedherein by reference in its entirety.

BACKGROUND OF THE INVENTION

Demand for wireless digital communication and data processing systems ison the rise. Inherent in most digital communication channels are errorsintroduced when transferring frames, packets or cells containing data.Such errors are often caused by electrical interference or thermalnoise. Data transmission error rates depend, in part, on the mediumwhich carries the data. Typical bit error rates for copper based datatransmission systems are in the order of 10⁻⁶. Optical fibers havetypical bit error rates of 10⁻⁹ or less. Wireless transmission systems,on the other hand, may have error rates of 10⁻³ or higher. Therelatively high bit error rates of wireless transmission systems posecertain difficulties in encoding and decoding of data transmitted viasuch systems. Partly because of its mathematical tractability and partlybecause of its application to a broad class of physical communicationchannels, the additive white Gaussian noise (AWGN) model is often usedto characterize the noise in most communication channels.

Data is often encoded at the transmitter, in a controlled manner, toinclude redundancy. The redundancy is subsequently used by the receiverto overcome the noise and interference introduced in the data whilebeing transmitted through the channel. For example, the transmittermight encode k bits with n bits where n is greater than k, according tosome coding scheme. The amount of redundancy introduced by the encodingof the data is determined by the ratio n/k, the inverse of which isreferred to as the code rate.

In a multiple-input multiple-output (MIMO) system, the transmitterincludes multiple transmit antennas and the receiver includes multiplereceive antennas. A MIMO system is typically used to increase the datarate, diversity, or a combination thereof. The increase in data rate isachieved by transmitting multiple data streams via the multiple transmitantennas, also known as spatial multiplexing. The diversity is achievedby increasing the redundancy between the transmit antennas through jointcoding.

The IEEE 802.11a standard defines data rates of 6 Mbps (megabits persecond) up to 54 Mbps. For some applications, higher data rates forgiven modulations and data rates higher than 54 Mbps are desirable.Other extensions, such as the use of MIMO systems and other extensionsmight be desirable. In order to avoid conflicts with existingstandardized communications and devices, extended devices that extendbeyond the limits of the 802.11a standard and legacy devices that complywith the existing standard and are not necessarily aware of extendedstandards both need to coexist in a common communication space andinteroperate at times.

Coexistence occurs where different devices can operate in a common spaceand perform most of their functions. For example, an extendedtransmitter transmitting to an extended receiver might coexist with alegacy transmitter transmitting to a legacy receiver, and the extendeddevices can communicate while the legacy devices communicate, or atleast where the two domains are such that one defers to the other whenthe other is communicating. Coexistence is important so that theadoption and/or use of extended devices (i.e., devices that are outside,beyond or noncompliant with one or more standards with which legacydevices adhere and expect other devices to adhere) do not requirereplacement or disabling of existing infrastructures of legacy devices.

Interoperability occurs where an extended device and a legacy device cancommunicate. For example, an extended transmitter might initiate atransmission in such a manner that a legacy device can receive the datasent by the extended transmitter and/or indicate that it is a legacydevice so that the extended transmitter can adjust its operationsaccordingly. For example, the extended transmitter might revert tostandards compliant communications or switch to a mode that, while notfully standards compliant, is available to the legacy receiver. Inanother situation, an extended receiver might successfully receive datafrom a legacy transmitter.

The IEEE 802.11a standard defines a 20 microsecond long preamble with astructure as shown in FIG. 1, having short training symbols S (0.8microseconds each), a guard interval LG, long training symbols L (3.2microseconds each) and a signal field (4 microseconds). The preamble isfollowed by data. The first eight microseconds include ten identicalshort training symbols that are used for packet detection, automaticgain control and coarse frequency estimation. The second eightmicroseconds include two identical long training symbols, L, preceded bya guard interval LG that is the same pattern as the last half (1.6microseconds) of the long training symbol L. The long training symbolscan be used for channel estimation, timing, and fine frequencyestimation.

FIG. 2 shows a long training sequence, L₁, that is used to generate thesignal representing the long training symbol in a conventional 802.11apreamble. This sequence represents values used over a plurality ofsubcarriers. As specified in the standard, the subcarriers span a 20 MHzchannel and with 64 subcarriers, they are spaced apart by 312.5 kHz. Byconvention, used here, the first value in the sequence is the value forthe DC subcarrier, followed by the value for the 1×312.5 kHz subcarrier,then the value for the 2×312.5=625 kHz subcarrier, etc., up to the 32ndvalue for the 31×312.5 kHz=9687.5 kHz subcarrier. The 33rd valuecorresponds to the −10 MHz subcarrier, followed by the −(10 MHz−312.5kHz) subcarrier, and so on, with the 64 value being for the −312.5 kHzsubcarrier. As can be seen from FIG. 1, the DC value and the 28ththrough 38th values, corresponding to the edges of the 20 MHz channel,are zero.

One approach to obtaining higher data rates is the use of morebandwidth. Another approach, used by itself or as well as the use ofmore bandwidth, is MIMO channels, where a plurality of transmitterstransmit different data or the same data separated by space to result inpossibly different multi-path reflection characteristics. When usingMIMOs or MISOs, a number of advantages are gained by detecting thenumber of transmit antennas at the receiver.

BRIEF SUMMARY OF THE INVENTION

In accordance with one embodiment of the present invention, each of anumber of preambles is adapted to be used in packets sent over awireless network, such as an 802.11a compliant wireless network, toenable detection of the number of transmit antennas. Packets containingpreambles of the present invention may be received by extended devicesas well as by legacy receivers that are not configured to receive andinterpret these preambles. The detection of the number of transmitantennas provides a number of advantages, such as enabling theestimation of the MIMO channel, synchronization of various transmittersand/or receivers, management of the application of power to variouscomponents disposed in the transmitters/receivers, and allocation ofmemory as well as other resource between various components of thereceiver/transmitters.

In one embodiment, extended-long training (ELT) symbols disposed in thepreamble are cyclically-shifted ELT symbols and transmitted on differenttransmit antennas. At the receiver (RX), if the received symbols aredetected as matching the known ELT symbols, a sum of shifted channelimpulse responses corresponding to the channels from the different TXantennas is attained. The time shift associated with the channel impulseresponse at the receiver indicates the number of times the ELT symbolsare cyclically shifted and thus represent the number of transmitantennas. In other words, by detecting the number of shifts; in thechannel impulse response at the receiver, the number of TX antennas isestimated. The ELT symbols may be selected to be the same as the 802.11along training symbols, but other long training symbols such as longtraining symbols where, next to the 802.11a subcarriers, the out-of-bandtones, i.e., the 28th through 38th subcarriers are disposed, may beused.

To detect the number of transmit antennas at the receiver, a fastFourier transform (FFT) operation is performed on the received samplesof the transmitted ELT symbols. Next, each subcarrier y_(q)(i) ismultiplied with the associated subcarrier frequency-domainrepresentation of the ELT symbol with a cyclic shift of zero so as toremove the effect of the ELT and obtain a frequency-domainrepresentation of the sum of shifted impulse responses. Next, inverseFourier transform or least squares (LS) operation is performed on themultiplied values to compute the transmission channel impulse response.The LS operation may be employed when fewer than all the tone of the ELTare used. The channel impulse response includes a sum of cyclicallyshifted impulse responses corresponding to the multiple transmitantennas. The number of cyclically shifted impulse responses in thechannel impulse response represents the number of transmit antennas.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows the preamble of a conventional 802.11a standard, as knownin the prior art.

FIG. 2 shows the frequency domain symbols of a long training symbolsequence starting with the DC subcarrier used in the 802.11a preamble ofFIG. 1, as known in the prior art.

FIG. 3 shows a number of devices coupled via a wireless network.

FIG. 4A is a plot of the absolute values of the channel impulse responseof a two-transmitter system, determined in accordance with oneembodiment of the present invention.

FIG. 4B is a plot of the real parts of a window integration of thepowers of the values used in FIG. 4A, in accordance with one embodimentof the present invention.

FIGS. 5A, 5B and 5C show modified preambles with cyclic shifts adaptedto be used by a receiver of a communication system having a multitude oftransmit antenna systems to detect the number of transmitters of such asystem, in accordance with one embodiment of the present invention.

FIG. 6 is a flow-chart of steps used to detect the number of transmitantennas of a communication system having a multitude of transmitantennas, in accordance with one embodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

In accordance with one embodiment of the present invention, each of anumber of preambles is adapted to be used in packets sent over awireless network, such as an 802.11a compliant wireless network, toenable detection of the number of transmit antennas. Packets containingpreambles of the present invention may be received by extended devicesas well as by legacy receivers that are not configured to receive andinterpret these preambles. The detection of the number of transmitantennas provides a number of advantages, such as enabling theestimation of the MIMO channel, synchronization of various transmittersand/or receivers, management of the application of power to variouscomponents disposed in the transmitters/receivers, and allocation ofmemory as well as other resource between various components of thereceivers/transmitters. The following description is provided withreference to MIMO systems, however, it is understood that the inventionequally applies to the multiple-input single-output (MISO) systems.

In accordance with some embodiments of the present invention,extended-long training (ELT) symbols disposed in the preamble arecyclically-shifted ELT symbols and transmitted on different transmitantennas. At the receiver (RX), if the received symbols are detected asmatching the known ELT symbols, a sum of shifted channel impulseresponses corresponding to the channels from the different TX antennasis attained. The time shift associated with the channel impulse responseat the receiver indicates the number of times the ELT symbols arecyclically shifted and thus represent the number of transmit antennas.In other words, by detecting the number of shifts in the channel impulseresponse at the receiver, the number of TX antennas is estimated.

FIG. 3 shows an exemplary wireless network used for communications amongtransmitters and receivers as indicated. As shown, two wireless devices1021, 1022 might use and interpret the modified preambles, while alegacy wireless device 104 might not be expecting the modifiedpreambles, but might hear signals representing such preambles. Extendedwireless devices 102 might operate using multiple channels and/ormultiple transmit antennas and/or multiple receive antennas. Whileseparate transmit and receive antennas are shown, antennas might be usedfor both transmitting and receiving in some devices.

Border 106 is not a physical border, but is shown to represent a spacewithin which signals can be received from devices within the space.Thus, as one device transmits a signal representing a packet withinborder 106, other devices within border 106 pick up the signals and, asthey are programmed, will attempt to determine if the signals representpackets and if so, then demodulate/decode the packets to obtain thetransmitted data.

The algorithm adapted to detect the number of transmit antennas isdescribed below with reference to a preamble with ELT symbols that arecyclically-shifted and transmitted from each of the transmit antennas ofthe MIMO system. In other words, the ELT symbols of the preamble arefirst transmitted from a first one of the transmit antennas and arethereafter cyclically-shifted the same number as the number of remainingtransmit antennas of the MIMO system and subsequently transmitted fromeach such transmit antenna. It is understood, however, that thealgorithm may also be applied to other training symbols.

To detect the number of transmit antennas at the receiver, a fastFourier transform (FFT) operation is performed on the received samplesof the transmitted ELT symbols. Assume r_(q)(n) represents the n-threceived time-domain sample on the q-th receive antenna. Further assumethat the FFT window of the training symbols includes N_(c) samples, andthe first sample of this FFT window corresponds to n=0. Applying the FFTyields the following equation (1):

$\begin{matrix}{{y_{q}(i)} = {C_{1}{\sum\limits_{n = 0}^{N_{c} - 1}{{r_{q}(n)}{\exp\left( {{–j}\; 2\pi\;\frac{i\; n}{N_{c}}} \right)}}}}} & (1)\end{matrix}$where y_(q)(i) represents the received information on receive antenna qand subcarrier i, and C₁ is a normalization constant.

Next, each subcarrier y_(q)(i) is multiplied with the associatedsubcarrier frequency-domain representation of the ELT symbol so as toremove the effect of the ELT and to maintain the channel information.Assume x_(p)(i) is the frequency-domain representation of the trainingsymbol on subcarrier i and transmit antenna p (or spatial stream p). Inthe following, it is assumed that a direct mapping of N_(s) spatialstreams to N_(t) transmit antennas occurs, therefore N_(s)=N_(t),although it is understood that the present invention may be readilyapplied to more general space-time-frequency mappings where the N_(s)spatial streams are not directly mapped to the N_(t) transmit antennas.The training symbol is assumed to be a known reference to both thetransmitter and receiver. Moreover, without loss of generality, it isassumed that no cyclic shift is applied to transmit antenna 1.Accordingly, the received frequency-domain information is multiplied bythe conjugate of x₁(i) to obtain the following:y′ _(q)(i)=y _(q)(i)x ₁*(i)  (2)where f(*) denotes the complex conjugation operation. Next, without lossof generality it is assumed that x_(p)(i) utilizes all subcarriers.Hence, an inverse FFT (IFFT) can be performed to transform the frequencydomain values of equation (2) into time domain values, as shown below:

$\begin{matrix}{{r_{q}^{\prime}(n)} = {C_{2}{\sum\limits_{i = 0}^{N_{c} - 1}{{y_{q}^{\prime}(i)}{\exp\left( {j\; 2\pi\;\frac{i\; n}{N_{c}}} \right)}}}}} & (3)\end{matrix}$where C₂ is a normalization constant.

Neglecting contributions due to noise, equation (3) may be rewritten asshown below:

$\begin{matrix}\begin{matrix}{{r_{q}^{\prime}(n)} = {C_{2}{\sum\limits_{i = 0}^{N_{c} - 1}{{y_{q}^{\prime}(i)}{\exp\left( {j\; 2\pi\frac{i\; n}{N_{c}}} \right)}}}}} \\{= {C_{2}{\sum\limits_{i = 0}^{N_{c} - 1}{\left( {\sum\limits_{p = 1}^{N_{t}}{{h_{f\;\_\; q\; p}(i)}{x_{p}(i)}}} \right){x_{1}^{*}(i)}{\exp\left( {j\; 2\pi\;\frac{i\; n}{N_{c}}} \right)}}}}} \\{{= {C_{3}{\sum\limits_{p = 1}^{N_{t}}{h_{t\;\_\;{qp}}\left( {n - s_{p}} \right)}}}},}\end{matrix} & (4)\end{matrix}$where h_(t) _(—) _(qp)(n) is the time-domain representation of thecommunication channel between transmitter p and receiver q, in otherwords, the impulse response of the communication channel betweentransmitter p and receiver q, and sp denotes the cyclic shift in samplesapplied to the p-th transmit antenna. C₃ is a normalization constant.

The above computations determine the degree of time-shift, if any, ofthe channel impulse response as detected by the receiver. In order toimprove the accuracy of such a detection, cyclic window integration maybe applied. Assume w(n) represents a window of N_(w) samples long.Accordingly, a cyclic window integration over the power of r′_(q)(n),i.e., |r′_(q)(n)|², with n=0, 1, . . . N_(c)−1, yields the following:

$\begin{matrix}{{a(n)} = {\sum\limits_{k = 0}^{N_{w} - 1}{{{r_{q}^{\prime}\left( {\left( {n + k} \right){mod}\mspace{11mu} N_{c}} \right)}}^{2}{w\left( {N_{w} - 1 - k} \right)}}}} & (5)\end{matrix}$where mod is the modulo operator. It is understood that instead of usingpower of r′_(q)(n) in the above equation, one may use, for example, theamplitude of r′_(q)(n) or any other suitable measure. The maximum valuesmeeting defined criteria in the intervals of expression (5) indicate thepresence or absence of impulse responses in these intervals. Theintervals are selected so as to match the various expected cyclic shiftsof the ELT symbols.

The following is an exemplary pseudo-code adapted to detect the numberof transmit antennas of a MIMO system by determining the presence ofmaximum values in the various intervals associated with the cyclicintegration windows, as defined in expression (5) above. The exemplarypseudo-code below assumes that the MIMO system includes no more thanfour transmit antennas. It is understood, however, that the followingpseudo-code may be readily modified to detect the number of transmitantennas in a MIMO system having any number of transmit antennas. Assumethat {M1, M2, . . . , M6} represent the maximum values in each ofintervals {I1, I2, . . . , I6} as shown in FIG. 4B for this example.Accordingly, the number of transmit antennas is detected as shown below:

 (1) {{if ((M2 > THR1) AND (M3 > THR1) AND (M4 > THR1)) N_(t) = 4  (2)elseif ((M5 > THR1) AND (M6 > THR1)) N_(t) = 3  (3) elseif (M3 > THR1)N_(t) = 2  (4) else N_(t) = 1  (5) if ((M2 > THR2) OR (M4 > THR2)) N_(t)= 4  (6) elseif ((M5 > THR2) OR (M6 > THR2)) N_(t) = 3  (7) if (mean(M2,M3, M4) > mean(M5, M6)) N_(t) = 4  (8) elseif (mean(M5,M6) > M3) N_(t) =3  (9) if (max(M1,M2,M3,M4) < THR3*min(M1,M2,M3,M4)) N_(t) = 4 (10)elseif (max(M1,M5,M6) < THR3*min(M1,M5,M6)) N_(t) = 3 (11) elseif(max(M1,M3) < THR3*min(M1,M3)) N_(t) = 2 (12) else N_(t) = 1}}

In the above pseudo-code, parameters THR1, THR2 and THR3 representadjustable threshold values that may vary with the noise power. Anestimate of the noise power may be obtained by subtracting the receivedsamples corresponding to two (or more) subsequent equivalent trainingsymbols and calculating the power of the result.

Lines 1-4 of the above pseudo-code, defining the first comparison rule,determine the maximum values that are above the designated thresholdvalues(s) or noise floors. Lines 5-6, defining the second comparisonrule, are used to verify whether certain peaks are above a higherthreshold, thus ensuring that even if the peak (maximum) values arebelow the noise floor, the correct number of transmit antennas isdetected. Lines 7-8 are used to distinguish between three TX and four TXcase because for high delay spread cases in the event the first twocomparison rules fail. Lines 9-12, defining the fourth comparison rule,are used to ensure that the highest and the lowest peak values arewithin a certain range (e.g., 15 or 20 dB). This is particularlyimportant to distinguish between the one TX and two TX cases.

Depending on the number of TX antennas, the result of the abovecomputations contains a sum of time-shifted impulse responses. FIG. 4Ashows the result of such computations when N_(t)=2, N_(c)=64, s₁=0 ands₂=32. In other words, in the computations associated with data shown inFIG. 4A, the cyclic shift of the ELT symbol at the first TX antenna is 0samples and that at the second TX antenna is 32 samples at a samplingrate of 20 MHz or, equivalently, 1.6 μs. When there are three transmitantennas, the shifts may be, e.g., 0, 21, and 42 samples; when there arefour transmit antennas, the shifts may be, e.g., 0, 16, 32, and 48samples, respectively, as understood by those skilled in the art. Theabsolute values of the complex numbers of the two TX channel impulseresponse is used in the plot of FIG. 4B.

In accordance with equation (5), for a window of 4 samples, integratingthe square powers of the values shown in FIG. 4A results in the realparts as shown in FIG. 4B. Therefore, as described above, equation (5)may be used after equation (4) to improve the accuracy of the detection.For an interval size of 4 samples, the intervals for all possible shiftsas mentioned in the exemplary pseudo code above are {I1, I2, I3, I4, I5,I6}={0-3, 16-19, 21-24, 32-35, 42-45, 48-51}, as shown in FIG. 4B.

FIGS. 5A, 5B and 5C respectively show cyclically shifted preamblesadapted for transmission from systems having respectively 2, 3 and 4transmit antennas. As used herein, a sequence in the frequency domain isexpressed with uppercase letters (e.g., L(i)), while the correspondingtime sequence is expressed with lowercase letters (e.g., l(i)).

The preambles which enable or enhance coexistence of MIMO packets inlegacy devices include a cyclic delay shift applied to the ELT as wellas Signal field prior to applying the guard time extension. For example,assume L(i) and D(i) are the 64 subcarrier values for the ELT and Signalfield symbols, respectively. For a conventional 802.11a singletransmitter transmission, the time samples for the long training symbolare derived by taking the 64-point IFFT of L(i) to obtain l(n) andtransmitting the samples of l(n). Thus, with the guard time, the ELTsymbol and guard time are constructed as [l(32:63) l(0:63) l(0:63)],i.e., the IFFT output is repeated twice and the last 32 samples areprefixed to form the long training guard interval. As with theconventional timing, the long training guard interval (32 samples) istwice as long as the guard interval for 802.11a data symbols (16samples). The signal field is formed by [d(48:63) d(0:63)], whered(0:63) are the 64 samples of the IFFT of D(i).

In the case of an exemplary two transmitter MIMO device, the firsttransmitter transmits the long training symbol and signal field, as isthe case with an of 802.11a transmission. The second transmitter,however, applies a cyclic shift such that instead of the IFFT outputl(0:63), the cyclically shifted samples ls=[l(32:63) l(0:31)] are usedto construct the long training symbol samples [ls(32:63) ls(0:63)ls(0:63)]. With respect to the signal field, the shifted samplesds=[d(32:63) d(0:31)] are used to construct the signal field as[ds(48:63) ds(0:63)].

In a legacy 802.11a packet, one 3.2 microsecond repetition of the longtraining symbol L as shown in FIG. 1 is expressed in the time domain asthe IFFT of L(i), where L(i) contains 64 subcarrier values, of which 52are non-zero. The time samples l(n) are given as shown in equation (6):

$\begin{matrix}{{l(n)} = {\sum\limits_{i = 0}^{63}{{L(i)}{\exp\left( {j\;\frac{2\pi\; i\; n}{64}} \right)}}}} & (6)\end{matrix}$

In accordance with the preambles adapted for detection of the number oftransmit antennas as well as for extended modes operations, L(i) maycontain more than 52 non-zero subcarriers. Furthermore, in the case ofMIMO transmission, l(n) may have a cyclic shift that may be differentfor each transmitter. The shifted signal l_(k)(n) can be derived froml(n) as l_(k)(n)=l([n+64−s_(k)]% 64), where “%” denotes the modulooperator and s_(k) is the cyclic delay of transmitter k in 20 MHzsamples. This expression assumes a 20 MHz sampling rate, such that thereare 64 samples in a 3.2 microsecond interval. An alternative method ofgenerating the cyclic shift is to apply a phase ramp rotation to allsubcarrier values of L(i) prior to calculating the IFFT, such as thatshown in equation (7) below:

$\begin{matrix}{{l_{k}(n)} = {\sum\limits_{i = 0}^{63}{{L(i)}{\exp\left( {{- j}\;\frac{2\pi\; i\; s_{k}}{64}} \right)}{\exp\left( {j\;\frac{2\pi\; i\; n}{64}} \right)}}}} & (7)\end{matrix}$

For a MIMO system with two transmit antennas and two different transmitdata streams, cyclic delay values s_(k) may be 0 and 32 samples,respectively, corresponding to a cyclic delay of 1.6 microsecondsbetween the two transmitters. For three transmitters, s_(k) may be 0,21, and 42 samples, respectively. For four transmitters, s_(k) may be 0,16, 32, and 48 samples, respectively.

FIG. 6 shows a flowchart 600 of steps carried out to detect the numberof transmit antennas. The process starts at step 602. At step 604, thetransmitted ELT symbols are received. At step 606 an FFT operation isperformed on the received samples of the transmitted ELT symbols,possibly averaged over the two consecutive ELT symbols. At step 606 theFFT values are multiplied with known frequency-domain representation ofthe training symbols on each sub-carrier of the reference transmitantenna, herein assumed transmit antenna 1. At step 608, an inverse FFTor LS estimate is performed on the multiplied results to transform thefrequency domain values of equation (2) into time domain values. At step612, the number of shifted impulse responses in the channel impulseresponse are isolated to determine the number of transmit antennas. Theprocess ends at step 614.

The above embodiments of the present invention are illustrative and notlimiting. Various alternatives and equivalents are possible. Theinvention is not limited by the type of encoding, decoding, modulation,demodulation, equalization, filtering, etc., performed. The invention isnot limited to the number of transmit or receive antennas. The inventionis not limited by the rate used to transfer the data. The invention isnot limited by the type of integrated circuit in which the presentdisclosure may be disposed. Nor is the disclosure limited to anyspecific type of process technology, e.g., CMOS, Bipolar, or BICMOS thatmay be used to manufacture the present disclosure. Other additions,subtractions or modifications are obvious in view of the presentdisclosure and are intended to fall within the scope of the appendedclaims.

What is claimed is:
 1. A method, comprising: receiving samples oftraining symbols that are cyclically shifted with respect to one anotherand transmitted from a plurality of transmit antennas; performingFourier transform operations on the received samples of the trainingsymbols to generate frequency-domain values associated with the receivedsamples; multiplying the associated frequency-domain values of thereceived samples with known frequency-domain representations of thetraining symbols on each sub-carrier of a reference transmit antenna togenerate multiplied values; computing a transmission channel impulseresponse, wherein computing the transmission channel impulse responseincludes performing a least square operation on the multiplied values;and detecting a number of shifted impulse responses in the transmissionchannel impulse response, wherein the detected number of shifted impulseresponses represents a number of the transmit antennas.
 2. A method,comprising: receiving samples of training symbols that are cyclicallyshifted with respect to one another and transmitted from a plurality oftransmit antennas; performing Fourier transform operations on thereceived samples of the training symbols to generate frequency-domainvalues associated with the received samples; multiplying the associatedfrequency-domain values of the received samples with knownfrequency-domain representations of the training symbols on eachsub-carrier of a reference transmit antenna to generate multipliedvalues; computing a transmission channel impulse response, whereincomputing the transmission channel impulse response includes performingan inverse Fourier transform operation on the multiplied values; anddetecting a number of shifted impulse responses in the transmissionchannel impulse response, wherein the detected number of shifted impulseresponses represents a number of the transmit antennas.
 3. A method,comprising: receiving samples of training symbols that are cyclicallyshifted with respect to one another and transmitted from a plurality oftransmit antennas; performing Fourier transform operations on thereceived samples of the training symbols to generate frequency-domainvalues associated with the received samples; multiplying the associatedfrequency-domain values of the received samples with knownfrequency-domain representations of the training symbols on eachsub-carrier of a reference transmit antenna to generate multipliedvalues; computing a channel impulse response; detecting a number ofshifted impulse responses in the transmission channel impulse response,wherein the detected number of shifted impulse responses represents anumber of the transmit antennas; applying cyclic window integration onthe channel impulse response; and detecting a presence or absence of apeak value in each cyclic window associated with the integration todetect the number of shifted impulse responses in the channel impulseresponse.
 4. The method of claim 3, further comprising: establishing oneor more threshold values to define one or more matching rules to detecta presence or absence of a peak value in each cyclic window.
 5. Themethod of claim 4, wherein the one or more threshold values and the oneor more matching rules are variables.
 6. The method of claim 3, whereina frequency-domain value on a receive antenna q and a subcarrier i isdefined by:${y_{q}(i)} = {C_{1}{\sum\limits_{n = 0}^{N_{c} - 1}{{r_{q}(n)}{\exp\left( {{- j}\; 2\pi\;\frac{i\; n}{N_{c}}} \right)}}}}$wherein r_(q)(n) represents an n-th received time-domain sample on aq-th receive antenna, wherein a Fourier transform window of the trainingsymbols includes N_(c) samples, and a first sample of the Fouriertransform window corresponds to n=0, wherein C₁ is a normalizationconstant; and wherein computing the channel impulse response includesperforming a least square operation on the multiplied values.
 7. Themethod of claim 6, wherein each multiplied value is defined by:y′ _(q)(i)=y _(q)(i)x ₁*(i) wherein x*₁(i) represents a frequency-domainrepresentation on the subcarrier i of a training symbol without cyclicshift, and wherein x*₁(i) is a complex conjugate of x₁(i).
 8. The methodof claim 7, wherein a time-domain representation of the channel impulseresponse is further defined by:${r_{q}^{\prime}(n)} = {C_{2}{\sum\limits_{i = 0}^{N_{c} - 1}{{y_{q}^{\prime}(i)}{\exp\left( {{j2\pi}\;\frac{i\; n}{N_{c}}} \right)}}}}$wherein C₂ is a normalization constant.
 9. The method of claim 8,wherein the time-domain representation of the channel impulse responseis further defined by: $\begin{matrix}{{r_{q}^{\prime}(n)} = {C_{2}{\sum\limits_{i = 0}^{N_{c} - 1}{{y_{q}^{\prime}(i)}{\exp\left( {j\; 2\pi\;\frac{i\; n}{N_{c}}} \right)}}}}} \\{= {C_{2}{\sum\limits_{i = 0}^{N_{c} - 1}{\left( {\sum\limits_{p = 1}^{N_{t}}{{h_{f\;\_\;{qp}}(i)}{x_{p}(i)}}} \right){x_{1}^{*}(i)}{\exp\left( {j\; 2\pi\;\frac{i\; n}{N_{c}}} \right)}}}}} \\{{= {C_{3}{\sum\limits_{p = 1}^{N_{t}}{h_{t\;\_\;{qp}}\left( {n - s_{p}} \right)}}}},}\end{matrix}$ wherein h_(t) _(—) _(qp)(n) is a time-domainrepresentation of a communication channel between a transmitter p and areceiver q, and s_(p) represents a cyclic shift in samples applied to atraining symbol of a p-th transmit antenna or spatial stream, andwherein C₃ is a normalization constant.
 10. The method of claim 9,wherein the cyclic window integration is defined by${a(n)} = {\sum\limits_{k = 0}^{N_{w} - 1}{{{r_{q}^{\prime}\left( {\left( {n + k} \right){mod}\mspace{14mu} N_{c}} \right)}}^{2}{w\left( {N_{w} - 1 - k} \right)}}}$wherein mod is a modulo operator, and wherein w(n) represents a windowof N_(w) samples long.
 11. A receiver configured to: receive samples oftraining symbols that are cyclically shifted with respect to one anotherbefore being transmitted from a plurality of transmit antennas; performFourier transform operations on the received samples of the trainingsymbols to generate frequency-domain values associated with the receivedsamples; multiply the associated frequency-domain values of the receivedsamples with a frequency-domain representation of the training symbolson each sub-carrier of a reference transmit antenna to generatemultiplied values; compute a channel impulse response by performing aleast squares operation on the multiplied values; and detect a number ofshifts of the channel impulse response representative of a number of thetransmit antennas.
 12. A receiver configured to: receive samples oftraining symbols that are cyclically shifted with respect to one anotherbefore being transmitted from a plurality of transmit antennas; performFourier transform operations on the received samples of the trainingsymbols to generate frequency-domain values associated with the receivedsamples; multiply the associated frequency-domain values of the receivedsamples with a frequency-domain representation of the training symbolson each sub-carrier of a reference transmit antenna to generatemultiplied values; compute a channel impulse response, wherein computingthe channel impulse response includes performing an inverse Fouriertransform operation on the multiplied values; and detect a number ofshifts of the channel impulse response representative of a number of thetransmit antennas.
 13. A receiver configured to: receive samples oftraining symbols that are cyclically shifted with respect to one anotherbefore being transmitted from a plurality of transmit antennas; performFourier transform operations on the received samples of the trainingsymbols to generate frequency-domain values associated with the receivedsamples; multiply the associated frequency-domain values of the receivedsamples with a frequency-domain representation of the training symbolson each sub-carrier of a reference transmit antenna to generatemultiplied values; and compute a channel impulse response to detect anumber of shifts of the channel impulse response representative of anumber of the transmit antennas; wherein the receiver is furtherconfigured to apply cyclic window integration on the channel impulseresponse and to detect a presence or absence of a peak value in eachcyclic window to detect the number of shifted impulse responses in thechannel impulse response.
 14. The receiver of claim 13, wherein thereceiver is further configured to establish one or more threshold valuesto define one or more matching rules to detect the presence or absenceof a peak value.
 15. The receiver of claim 14, wherein the one or morethreshold values and the one or more matching rules are variables. 16.The receiver of claim 13, wherein a frequency-domain value on a receiveantenna q and a subcarrier i is defined by:${y_{q}(i)} = {C_{1}{\sum\limits_{n = 0}^{N_{c} - 1}{{r_{q\;}(n)}{\exp\left( {{- j}\; 2\pi\;\frac{i\; n}{N_{c}}} \right)}}}}$wherein r_(q)(n) represents an n-th received time-domain sample on aq-th receive antenna, wherein a Fourier transform window of the trainingsymbols includes N_(c) samples, and a first sample of the Fouriertransform window corresponds to n=0, wherein C₁ is a normalizationconstant, and wherein to compute the channel impulse response, thereceiver is further configured to perform a least square operation onthe multiplied values.
 17. The receiver of claim 16, wherein eachmultiplied value is defined by:y′ _(q)(i)=y _(q)(i)x ₁*(i) wherein x₁(i) represents a frequency-domainrepresentation on a subcarrier i of a training symbol without cyclicshift, and wherein x*₁(i) is a complex conjugate of x₁(i).
 18. Thereceiver of claim 17, wherein a time-domain representation of thechannel impulse response is further defined by:${r_{q}^{\prime}(n)} = {C_{2}{\sum\limits_{i = 0}^{N_{c} - 1}{{y_{q}^{\prime}(i)}{\exp\left( {j\; 2\pi\;\frac{i\; n}{N_{c}}} \right)}}}}$where C₂ is a normalization constant.
 19. The receiver of claim 18,wherein the time-domain representation of the channel impulse responseis further defined by: $\begin{matrix}{{r_{q}^{\prime}(n)} = {C_{2}{\sum\limits_{i = 0}^{N_{c} - 1}{{y_{q}^{\prime}(i)}{\exp\left( {j\; 2\pi\;\frac{i\; n}{N_{c}}} \right)}}}}} \\{= {C_{2}{\sum\limits_{i = 0}^{N_{c} - 1}{\left( {\sum\limits_{p = 1}^{N_{t}}{{h_{f\;\_\;{qp}}(i)}{x_{p}(i)}}} \right){x_{1}^{*}(i)}{\exp\left( {j\; 2\pi\;\frac{i\; n}{N_{c}}} \right)}}}}} \\{= {C_{3}{\sum\limits_{p = 1}^{N_{t}}{h_{t\;\_\;{qp}}\left( {n - s_{p}} \right)}}}}\end{matrix}$ wherein h_(t) _(—) _(qp)(n) is a time-domainrepresentation of the communication channel between a transmitter p anda receiver q, and s_(p) represents a cyclic shift in samples applied totraining symbol of a p-th transmit antenna or spatial stream, andwherein C₃ is a normalization constant.
 20. The receiver of claim 19,wherein a cyclic window integration is further defined by:${a(n)} = {\sum\limits_{k = 0}^{N_{w} - 1}{{{r_{q}^{\prime}\left( {\left( {n + k} \right){mod}\mspace{11mu} N_{c}} \right)}}^{2}{w\left( {N_{w} - 1 - k} \right)}}}$wherein mod is a modulo operator, and wherein w(n) represents a windowof N_(w) samples long.
 21. A receiver, comprising: means for receivingsamples of training symbols that are cyclically shifted with respect toone another before being transmitted from a plurality of transmitantennas; means for performing Fourier transform operations on thereceived samples of the training symbols to generate frequency-domainvalues associated with the received samples; means for multiplying theassociated frequency-domain values of the received samples with afrequency-domain representation of the training symbols on eachsub-carrier of a reference transmit antenna to generate multipliedvalues; means for computing a channel impulse response by performing aleast squares operation on the multiplied values; and means fordetecting a number of shifts of the channel impulse responserepresentative of a number of the transmit antennas.
 22. A receiver,comprising: means for receiving samples of training symbols that arecyclically shifted with respect to one another before being transmittedfrom a plurality of transmit antennas; means for performing Fouriertransform operations on the received samples of the training symbols togenerate frequency-domain values associated with the received samples;means for multiplying the associated frequency-domain values of thereceived samples with a frequency-domain representation of the trainingsymbols on each sub-carrier of a reference transmit antenna to generatemultiplied values; means for computing a channel impulse response,wherein computing the channel impulse response includes performing aninverse Fourier transform operation on the multiplied values; and meansfor detecting a number of shifts of the channel impulse responserepresentative of a number of the transmit antennas.
 23. A receiver,comprising: means for receiving samples of training symbols that arecyclically shifted with respect to one another and transmitted from aplurality of transmit antennas; means for performing Fourier transformoperations on the received samples of the training symbols to generatefrequency-domain values associated with the received samples; means formultiplying the associated frequency-domain values of the receivedsamples with known frequency-domain representations of the trainingsymbols on each sub-carrier of a reference transmit antenna to generatemultiplied values; means for computing a channel impulse response; meansfor detecting a number of shifted impulse responses in the transmissionchannel impulse response, wherein the detected number of shifted impulseresponses represents a number of the transmit antennas; means forapplying cyclic window integration on the channel impulse response; andmeans for detecting a presence or absence of a peak value in each cyclicwindow associated with the integration to detect the number of shiftedimpulse responses in the channel impulse response.